Systems and methods for blocking microwave propagation in parallel plate structures

ABSTRACT

Systems and methods are taught for blocking the propagation of electromagnetic waves in parallel-plate waveguide (PPW) structures. Periodic arrays of resonant vias are used to create broadband high frequency stop bands in the PPW, while permitting DC and low frequency waves to propagate. Some embodiments of resonant via arrays are mechanically balanced, which promotes improved manufacturability. Important applications include electromagnetic noise reduction in layered electronic devices such as circuit boards, ceramic modules, and semiconductor chips.

BACKGROUND

The present application is a divisional application of U.S. Ser. No.10/796,398, filed on Mar. 8, 2004 which issued as U.S. Pat. No.7,157,992, on Jan. 2, 2007, and is co-pending with U.S. Ser. No.11/374,931, filed on Mar. 13, 2006, which is a continuation of U.S. Ser.No. 10/796,398, each application being incorporated herein by reference.

1. Field of the Inventions

The field of the invention relates generally to systems and methods forblocking the propagation of electromagnetic waves in parallel platestructures and more particularly, to electromagnetic noise reduction inlayered electronic devices.

2. Background Information

Electronic devices are often configured as layered assemblies oftwo-dimensional structures. These layered assemblies, which include suchfamiliar devices as printed wiring boards, multi-chip modules, andintegrated semiconductor chips, are known generally as “multi-layerpreformed panel circuits” or simply “panel circuits”. Electromagneticnoise propagation in panel circuits has become increasingly problematicas increases in performance dictate simultaneously higher deviceoperating speeds and decreased operating voltages. Both of these trendshave converged to make these devices more susceptible to electronicnoise, thereby limiting panel circuit performance because of spuriouselectrical signal levels generated primarily within the panel circuititself. One particularly troublesome noise source is high-speed digitalswitching noise imposed on the nominally constant voltage (DC) powerdistribution system. DC power distribution is most commonly accomplishedin panel circuits by means of two closely spaced and substantiallyparallel conductors, generally referred to as the “power plane” and the“ground plane”. This general structure unfortunately also enables radiofrequency (RF) noise propagation throughout the device by acting as aparallel-plate waveguide (PPW). Various means have been employed toattenuate this electronic noise by, for example, inserting strategicallyplaced shunt capacitors and/or selecting the location of sensitivecomponents to correspond to voltage minima in the noise spatialdistribution. These methods are relatively effective at frequenciesbelow about 500 MHz. Above this frequency level, there remains a growingneed for more effective means of electrical noise isolation.

One possible approach to mitigating the effect of power plane noise inpanel devices would be to impose within the power plane, RF blockingfilter structures that could operate effectively above 500 MHz. Aparticularly effective design of high-frequency RF filter is the ‘waffleiron’ filter, first proposed in 1950 by S. B. Cohn. Waffle iron filtersare a type of low-pass corrugated waveguide filter that contains bothtransverse and longitudinal slots cut into the internal walls of therectangular guide. These slots create opposing teeth or bosses,resulting in the structure suggesting its name. Waffle iron waveguidefilters, as depicted generically in FIG. 1, are characterized by wide,high-attenuation, two-dimensional “stop bands”. Stop bands are spectralregions where electromagnetic wave propagation is impeded due to theconstraints imposed by a periodic array of interactive elements. Thetopology, capacitance, and inductance of the waffle iron filter'sresonant elements define its stop band characteristics. Waffle ironfilters were originally designed for high power microwave applications,and its embodiments for this application are not well suited tosmall-scale panel circuits. However, the technologies that have arisenin support of circuit board, integrated circuit, and other types ofpanel circuit fabrication are adaptable to making various types ofminiaturized resonant elements that could be configured to producecompact filters that have a similar effect, or characteristics of awaffle iron type of filter.

For example, in printed wiring boards (PWB's), “resonant vias” can befabricated. A resonant via is used here to denote a shunt electricalcircuit containing one or more plated through holes (PTH) in series withone or more capacitors. PTH's are routinely fabricated in PWB's, as wellas in other panel circuit devices such as multi-chip modules andintegrated ciruits (IC's). The term “resonant via” was introduced bySedki Riad in his U.S. Pat. No. 5,886,597 (Riad). However, Riad morenarrowly claimed a resonant via as one PTH in series with one capacitorand his patent emphasizes only the RF decoupling application forresonant vias, where they are used as a low impedance interlayerconnection between metal layers in multilayer PWB's. Although Riaddiscloses the use of multiple resonant vias, he does not suggest theidea of employing a periodic array for the purpose of creating a welldefined stop band filter.

SUMMARY OF THE INVENTION

A method for configuring systems of RF filters adaptable to panel typecircuits such as printed wiring boards, integrated circuits, multi-chipmodules and the like. In one aspect, a generalized method for generatinga comprehensive set of resonant via topologies and the incorporation ofthese topologies into systems of periodic arrays are adaptable to theproblem of reducing RF noise within parallel plate waveguide (PPW)structures. More particularly, the method disclosed address the problemof millimeter and microwave propagation within the parallel PPWstructures that are inherent in the generally layered character of panelcircuits.

These and other features, aspects, and embodiments of the invention aredescribed below in the section entitled “Detailed Description of thePreferred Embodiments.”

BRIEF DESCRIPTION OF THE DRAWINGS

Features, aspects, and embodiments of the inventions are described inconjunction with the attached drawings, in which:

FIG. 1 illustrates a prior art waffle-iron waveguide filter.

FIG. 2 illustrates an elevation view showing an array of resonant vias.

FIG. 3 illustrates an equivalent circuit for a resonant via shown inFIG. 2.

FIG. 4 illustrates five basic types of resonant vias.

FIG. 5 illustrates a wire periodically loaded with uniform seriesimpedances, an array of such loaded wires to create a loaded wiremedium, and a special case of this medium using parallel-platecapacitors as circuit loads.

FIG. 6 illustrates planes of symmetry in a capacitive loaded wiremedium, and several exemplary equivalent parallel-plate waveguide (PPW)stop band filters.

FIG. 7 illustrates an alternative implementation of a loaded wiremedium, and several embodiments of PPW stop band filters derived usingplanes of symmetry.

FIG. 8 illustrates hybrid embodiments of PPW stop band filters thatinclude both internal and external capacitive loads.

FIG. 9 illustrates an elevation view of an array of internal T resonantvias.

FIG. 10 illustrates the Brillouin Zone for a rectangular periodic arraywhose periods in the x and y directions are “a” and “b”.

FIG. 11 presents the dispersion diagram along the ΓX line for the squarelattice of resonant vias shown in FIG. 9.

FIG. 12 presents the dispersion diagram along the ΓM line for the squarelattice of resonant vias shown in FIG. 9.

FIG. 13 is the simulated geometry for the resonant via array shown inFIG. 9.

FIG. 14 presents the TEM mode transmission and reflection response forthe array of resonant vias shown in FIG. 13.

FIG. 15 presents a comparison of the stopband performance predicted bytwo methods: numerical (Microstripes) simulation on a finite sized arrayversus an eigenmode calculation of an infinite array.

FIG. 16 illustrates an equivalent loaded wire medium for a given arrayof internal T resonant vias, as well as the equivalent circuit for theprincipal axis wave propagation in the medium.

FIG. 17 illustrates various embodiments of resonant via arrays withcommensurate periods.

FIG. 18 illustrates a commensurate periodic array comprising a cascadeof eight unit cells of resonant vias of type H12 shown in FIG. 17.

FIG. 19 presents the TEM mode transmission and reflection response forthe cascaded commensurate array of resonant vias shown in FIG. 18.

FIG. 20 illustrates an embodiment of a PPW stop band filter integratedinto the power distribution network of a multilayered preformed panelcircuit.

FIG. 21 illustrates a resonant via comprising a plated through-hole asit might be realized in a printed wiring assembly (PWA).

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 2 depicts a general example of the invention of Riad, implementinga grounding scheme by a plurality of resonant vias interposed between aPPW structure as disclosed in FIG. 2 of Riad. FIG. 3 shows Riad'sequivalent circuit model of a single resonant via of FIG. 2, whichconsists of an inductor (LV) in series with a capacitor (CV) thatmodels, respectively, the inductive via post 14 a and capacitive plate14 b of FIG. 2. This idealized circuit comprises a resonant shuntcircuit between the upper plate 4 and lower ground plate 10. Riad showedthat by appropriately configuring the dimensions of the vias, particularvalues of inductance LV and capacitance CV are produced so that theimpedance of the shunt circuit can be made essentially zero in thedesired region around the resonant frequency and finite elsewhere.Riad's resonant vias have many applications for circuit boards, such asfor shunting spurious signals of a particular frequency.

Riad, does not suggest, however, the use of the structures disclosedtherein in periodic arrays, wherein the periodicity of the arraycombines with the resonant character of the vias to effect a stop bandRF filter for PPW modes. What is not taught, and which is disclosedhere, is the utility of combining the teachings of Riad for resonantvias with that pertaining to waffle iron filters. By combining theseteachings, certain RF filter topologies that are particularly adaptableto the problem of blocking RF noise in PWB's in particular, and panelcircuits in general, can be developed in accordance with the systems andmethods described below. There are currently known in the art threetopologies that are adaptable as resonant vias, the simple “T” andexternal “T” topologies disclosed by Riad, and the buried single layercapacitor internal double “T” disclosed in U.S. Pat. No. 6,542,352 toDevoe et. al. (Devoe). These three topologies are presented togetherwith three newly conceived resonant via topologies in FIG. 4. All thebasic topologies of FIG. 4 can be conveniently categorized as either “T”or “I”, and further categorized according to whether they are“internal”, “external”, or “hybrid” combinations thereof. The termhybrid is used to define configurations that include both internal andexternal capacitive pads or patches. The topologies can be additionallycategorized as to whether they are mechanically “balanced” or“unbalanced”, as will be elucidated below.

In addition to the specific new topologies advanced in FIG. 4, a methodis presented below for deriving a comprehensive collection of filtertopologies using the topologies included in FIG. 4. Specifically, acomprehensive set of resonant element topologies can be derived bystarting with a generalized array of periodically loaded conductorsaligned perpendicular to the PPW. This electromagnetic model has beenapplied, for example, by Sergei Tretyakov for analyzing man-mademetamaterials. Tretyakov's loaded wire media model provides a verygeneral and powerful tool for generating a wide assortment of resonantelement topologies when combined with the systems and methods describedherein.

FIG. 5 depicts an idealized loaded wire array model as suggestedTretyakov. It consists of an evenly distributed array of parallel wires,each of which consists of a periodic sequence of inductors andcapacitors. In this particular example, the inductors (wire segments)and the capacitors (parallel plate segments) are in registry along the Zdirection and are equally space in the X and Y dimensions, thus forminga three-dimensional lattice structure. The utility of this model in thepresent context is not readily apparent until it is realized, as taughtherein, that a plane of symmetry in X & Y can be replaced by aconducting surface. By truncating the loaded wire media with aconducting plane along planes of symmetry, the electrical properties ofthe periodic array are unchanged in the X & Y dimensions. The reason whythe periodic loaded wire media may be truncated with conductors in thismanner is that the planes of symmetry for the infinite (in the Zdirection) wire media are also electric walls, defined to be locationswhere the tangential electric field goes to zero. As will be presentlydemonstrated, this procedure provides a powerful method for deriving agreat many different, but electrically equivalent, topologies.

FIG. 6 a depicts a segment of the Tretyakov loaded wire media model thatincludes four periods of three wires of the loaded wire array of FIG. 5.Locations A and C, and B and D are at the planes of symmetry of thecapacitors and inductors, respectively. FIG. 6 b shows the correspondingstructures obtained by truncating the periodic loaded wires withconducting metal surfaces between various planes of symmetry. It isimportant to note that all of the structures denoted in FIG. 6 b areelectrically equivalent as stop band filters and have exactly the sameelectromagnetic wave propagation characteristics in the X & Y dimensionsas the original infinite three-dimensional structure.

This same essential methodology can be extended then to include periodicloaded wires where the loads are staggered along the Z direction as ispresented in FIG. 7 a. Shown in FIG. 7 b is an exemplary embodimentobtained by truncating along the corresponding planes of symmetry E andG in FIG. 7 a. This method for generating equivalent topologies can beextended even further by recognizing that the capacitor plates can belocated external to the conducting planes. Examples of external varianttopologies are shown in FIG. 7 c. As can be appreciated, due to thelonger via length, the PTH wire diameter must be increased in theexternal variant to maintain the same total inductance as its internalequivalent. Topologies that are even more complex can be generated bycombining two or more of the topologies obtained by the symmetry planemethod, including various hybrid combinations of internally andexternally located capacitor plates.

FIG. 8, for example, presents six possible hybrid topologies. Theimpedances of all the individual resonant elements, external andinternal, can be configured to be equal, or periodic portions of eachcan be configured to have different values, thus providing the engineerwith useful additional design freedom with regard to the RF filtercharacteristics.

As can be appreciated from the above description, a virtually unlimitednumber of topologies can be generated by starting with any one of themany conceivable three-dimensional periodic loaded wire latticestructures and then applying the symmetry plane methodology describedabove. For example, topologies that are substantially symmetric about aplane midway between the parallel plates of the PPW are particularlyuseful for the construction of printed wiring boards because they areless prone to warping under thermal stress. Topologies that aresubstantially free from warping under thermal stress are considered“mechanically balanced”. It is not necessary for the topology to beexactly symmetric in order to effect a mechanically balanced structure.Which structures are, and are not, sufficiently symmetric to bemechanically balanced will be influenced by a particular implementationor technology.

A mechanically balanced panel circuit can be defined as containing (1)an even number of metal layers, and (2) a plane of symmetry with respectto the dielectric cores; i.e., the core thicknesses and materialproperties including coefficient of thermal expansion are mirrored aboutthe plane of symmetry. Any asymmetric topology can be converted to anequivalent symmetric structure by simply doubling its periodic spacingand then superimposing upon it the same topology inverted, i.e., rotated180 degrees, and shifted by a half a period. For example, in FIG. 6 b,AB_(int) and BC_(int) are inversions of the same imbalanced topology,and combining them as described results in a symmetric and mechanicallybalanced structure.

It is also recognized that the dimensions of the various topologies canbe adjusted in order to effect particular values of inductance andcapacitance, e.g., by varying the length and/or diameter of the wiresections to adjust inductance and the area and/or separation of thecapacitor plates (pads) to adjust capacitance. It will often beadvantageous to maximize the capacitance of the resonant via. Becausecapacitance increases in proportion to pad area, use of the areaavailable for the pads becomes important. FIG. 5 illustrates arectangular array of loaded wires, for which rectangular padsefficiently use the available area. However, other periodic arraylattices are possible for stop band filters, such as triangular orhexagonal arrays of wires, for which efficient pad shapes will have ahexagonal or triangular geometry, respectively. The dielectricproperties of the material in which the resonant vias are embedded canalso be advantageously selected to better meet certain goals, e.g., amaterial with high dielectric constant (permittivity) can be chosen toincrease the capacitance for a given plate spacing. These impedanceparameters, together with the spacing and periodicity of the resonantelements, can be used in concert to design filters with the desired stopband characteristics, as is known to those in the art familiar with thetheory of wave propagation in periodic structures.

The theory of wave propagation in periodic resonant structures, as firstpropounded by Cohn and further developed by others including Tretyakov,provides the design engineer with a solid basis for designing RF filterswith predictable behavior. Although one can employ various numerical andanalytical approaches to solving the filter problem, the treatmentprovided by Tretyakov for modeling loaded wire media will be applied byway of example.

Consider an example comprised of a square lattice of internal T resonantvias containing square pads and round vias of radius r. An elevationview of the array is shown in FIG. 9. Let the period be 400 mils in bothx and y directions such that a=b=400 mils. Let t1=4 mils and t2=22 milssuch that the entire thickness of the PPW is 26 mils assuming theconductors are modeled with zero thickness. Assume the blind vias have adiameter of 20 mils (radius r=10 mils) and a length t2 of 22 mils. Thesize of square pads is s=340 mils, hence a 60-mil gap exists between allpads. A homogeneous isotropic dielectric of relative permittivity∈_(r)=4.2 fills the entire height of the PPW. The first Brillouin zonefor a periodic structure with a rectangular lattice is shown in FIG. 10.The shaded boundary shows the domain in reciprocal space, or wave numberspace, for allowed values of wave numbers kx and ky. The ΓXM triangle iscalled the irreducible Brillouin zone because knowledge of thedispersion solution in this triangular domain will allow the entiredispersion surface in the rectangular domain to be identified throughsymmetry arguments.

The important thing to understand from the Brillouin Zone diagram isthat the propagation constants for wave propagation in the x directiononly, a principal axis where only kx is nonzero, can be found byexamining the dispersion diagram along the ΓX line. Propagationconstants for waves traveling at 45° with respect to the x and y axis(kx=ky) can be found by solving for the dispersion diagram along the ΓMline. Knowledge of the dispersion diagram along both of these domainswill reveal a complete stop band if it exists. Tretyakov's eigenvalueequation for loaded wire media, where the eigenwave propagationconstants for the x, y, and z directions are defined as q_(x), q_(y),and q _(z), can be rearranged into the more familiar form associatedwith a shunt loaded transmission line:

$\begin{matrix}{{\cos\left( {q_{x}a} \right)} = {{\cos\left( {k_{x}^{(0)}a} \right)} + {j\frac{\eta_{o}/\sqrt{ɛ_{r}}}{2\left( {Z_{s} + {\frac{k_{x}^{(0)}}{b}\frac{b}{c}Z_{load}}} \right)}{{\sin\left( {k_{x}^{(0)}a} \right)}.}}}} & (1)\end{matrix}$where η_(o) is the wave impedance of free space, Z_(s) is the shuntimpedance for a planar grid of wires so, and Z_(load)=2/(jωC) is theload impedance presented by the capacitive loads on the wires. Thecapacitance C=s²∈_(r)∈_(o)/t₁ is the parallel-plate capacitance betweenthe pads and the upper parallel plate as shown in FIG. 9. The parametersa, b, and c are the lattice constants, or the periods, in the x, y, andz directions respectively. The period c in the wire media can bedetermined using c=2(t₁+t₂). The propagation constant k_(x) ^((o)) isthe x component of the n=0 Floquet mode wave vector defined moregenerally by

$\begin{matrix}{k_{x}^{(n)} = {{- j}\sqrt{\left( {q_{y} + \frac{2n\;\pi}{b}} \right)^{2} + q_{z}^{2} - k^{2}}}} & (2)\end{matrix}$where k=ω√{square root over (μ_(o)∈_(o)∈_(r))} is the wave number forthe host dielectric medium. Tretyakov has also derived a formula for theshunt impedance for a wire grid:

$\begin{matrix}{Z_{s} = {j\frac{\eta}{2\sqrt{ɛ_{r}}}{\quad\left\lbrack {{\frac{{bk}_{x}^{(0)}}{\pi}{\ln\left( \frac{b}{2\pi\; r} \right)}} + {k_{x}^{(0)}\left\{ {{\sum\limits_{n \neq 0}{\frac{1}{k_{x}^{(n)}}\frac{\sin\left( {k_{x}^{(n)}a} \right)}{{\cos\left( {k_{x}^{(n)}a} \right)} - {\cos\left( {q_{x}a} \right)}}}} - \frac{b}{2\pi{n}}} \right\}}} \right\rbrack}}} & (3)\end{matrix}$where r is the wire radius (via radius). The series in equation (3) hasnegative and positive integer indices but omits the n=0 term. Itconverges very rapidly. The eigenvalue equation (1) can be solvednumerically for the eigenfrequencies as a function of the wave vector(q_(x), q_(y), q_(z)). Since propagation in a PPW is limited to wavestraveling in the lateral (XY) directions, then q_(z)=0. The solution isactually a set of surfaces in the domain of the Brillouin zone. Theeigenvalue equation (1) is real valued for reactive loads, so thesolutions can be found using straightforward numerical root findingtechniques.

Next, the eigenfrequencies are calculated along the domain of lines ΓXand ΓM. FIG. 11 shows the eigenfrequency solution, known as a dispersiondiagram, for the ΓX line. This is for TEM mode wave propagation along aprincipal axis, either the X or Y-axis. The line 1101, is the lightline, which defines the wavenumbers for a plane wave traveling throughthe host dielectric medium. The slope of the line 1101 is the phasevelocity for a TEM mode in the PPW for the case where no resonant viasare present. The lines 1102-1107 are the eigenfrequency solution to theeigenvalue equation (1). Note that there exist three well-defined stopbands below 20 GHz.

FIG. 12 shows the dispersion diagram for the ΓM line, which is for wavepropagation in the inter-cardinal direction 45° from either principalaxis (x and y axis). In this diagram, q_(x)=q_(y) and 0≦q_(x)≦π/a. Below20 GHz there exists two stop bands, but only the fundamental stop bandcoincides in both dispersion diagrams. Common to wave propagation inboth directions is a stop band extending from 1.505 GHz to 4.036 GHz. Sothis frequency range can be viewed as the electromagnetic band gap (EBG)for the capacitive-loaded wire media. In the equivalent PPW filter ofFIG. 9, TEM waves traveling in any lateral (XY) direction will be cutoff over this frequency range; i.e., they are evanescent waves.

To validate some of the numerical solutions from the eigenvalueequation, a full-wave electromagnetic simulation was performed usingMicrostripes™, a computational software tool available from Flomerics,Inc in Southborough, Mass. The simulated geometry is shown in FIG. 13.This is a cascade of eight unit cells along the x-axis. In thissimulation, the dielectric core has a relative permittivity of 4.2 tosimulate the FR4 material commonly used for fabricating PWB's. Forsimplicity, no dielectric loss is included, all metals are modeled aslossless, and all conductors have zero thickness. This PPW filter wasexcited as a two port with TEM mode ports defined approximately one unitcell from each end of the cascaded unit cells. The boundary conditionsat the sides of the grid (y_(min) and y_(max)) are magnetic walls due tothe symmetry of the unit cells. Note that for this direction ofpropagation (+X-axis) there is a magnetic wall through the center ofeach unit cell and along the sides.

The simulated transmission response for TEM waves 1401 is shown in FIG.14. At least three stop bands are clearly identified. Using the −10 dBlevel as a gauge for defining the band edges for the stop bands, asummary of the stop band frequencies is shown by the table in FIG. 15.Excellent agreement is seen between the eigenvalue solution and thefull-wave simulation for the comparison case of propagation in the xdirection. Hence, the equivalence between a loaded wire media and anarray of resonant vias is demonstrated numerically.

The simplest possible model is often helpful for gaining insight.Circuit models are generally the simplest and most powerful models forengineers to use given the availability of modern software tools. It is,therefore, useful to develop an equivalent circuit model for the PPWstop band filter to which these software analysis tools can be applied.Consider FIG. 16, which illustrates an array of internal T resonant viasin part (a), its equivalent loaded wire media in part (b), and anequivalent circuit model for the wire media in part (c). The inductanceof a wire of length c in the wire media is given by

$\frac{Z_{s}}{j\omega}\frac{c}{b}$where Z_(s) is given by equation (3) above. This inductance is frequencydependent, and it actually exhibits a resonance just below the thirdstop band. However, at low frequencies, such as near the fundamentalstop band, this inductance can be approximated with the very simplefrequency independent equation shown in FIG. 16( c). The capacitance Cis simply the parallel-plate capacitance between the pad and the upperplate. The characteristic impedance Z_(o) is defined by the height c andwidth b of one unit cell in the infinite wire media. One unit cell ofthe infinite wire media has an equivalent circuit for TEM modepropagation consisting of a transmission line with a shunt LC load. Thefundamental stop band can be readily determined from this model by usingstandard techniques such as a calculation of the image propagationconstant or image impedance. Topology type AB_(int) has been used inthis derivation. However, simple transmission line models areappropriate for all of the PPW stop band filter types derived from FIG.5( b) or 7(a).

It can be advantageous in terms of the resulting stop bandcharacteristics to combine in one overall structure, arrays having morethan a single period. This can be readily accomplished by superimposingon the base periodic structure, one or more “commensurate” periodicstructures, which are integer multiples of the base period. In otherwords, a commensurate array has multiple repeated features within alarger unit cell. Some examples of the many possible embodiments ofcommensurate period resonant via arrays are illustrated in FIG. 17. Allof the examples in FIG. 17 are linear combinations of resonant via typesthat can be derived using the loaded wire media model as describedabove. Most of these examples employ scaling in the lateral dimensionsas well as dual period translations to achieve the final structure. Asimple example is array E1 where larger external pads are arrayed in acheckerboard fashion and sets of four smaller external pads fill in theareas between the larger pads.

The merit of arrays with commensurate periods is that thesesuper-lattice filters can be shown to offer much broader stop bandperformance than single period filters. Consider the internal array H12in FIG. 17. A Microstripes™ simulation was run to calculate RFtransmission through a cascade of eight H12 unit cells. The simulatedstructure, shown in FIG. 18, has a period of 400 mils defined by thelarger pads. Large pads are 280 mils square and small pads are 180 milssquare. The small pads have a 200 mil period, or one-half that of thelarger pads. All vias are 20 mils square and either 22 or 30 mils intotal length corresponding to internal or external vias, respectively.The upper parallel plate of the PPW has apertures 40 mils square toallow the vias connecting the exterior pads to penetrate withouttouching. All pads are spaced 4 mils from the PPW plates, giving thestructure a total thickness of 30 mils. The dielectric core has arelative permittivity (dielectric constant) of 4.2 to simulate FR4. Thisdielectric is assumed to have a loss tangent of 0.02 at 4 GHz. Allmetals are modeled as lossless, and all conductors have zero thickness.This PPW filter was excited as a two port with TEM mode ports definedapproximately one unit cell from each end of the eight-cascaded cells.The boundary conditions at the sides of the grid (Y_(min) and Y_(max))are magnetic walls due to the symmetry of the unit cells. Note that forthis direction of propagation (+X axis) there is a magnetic wall throughthe center of each unit cell and along the sides.

The transmission response 1901 is shown in FIG. 19. Observe that afairly wide fundamental stop band exists from about 1.33 GHz to 10.0GHz, as defined by the −10 dB level on the S21 plot. Note that thiscommensurate array PPW filter has the same period as the simplerresonant via array of FIG. 13. The merit of this more complex array ofresonant vias can be seen by comparing the simulation results with thatof FIG. 14. It can be seen that incorporating an additional commensuratearray has moved the upper edge of the fundamental stop band up from 3.9GHz to 9.67 GHz, so the fundamental stop band now covers a frequencyratio of about 7.5:1 compared with 2.8:1 for the simple periodic array.This particular simulation is only an example to demonstrate the generalstop band broadening effect that is achievable with commensurate arraysand has not been optimized for any particular filter characteristic.

FIG. 20 shows the conceptual layout of a typical eight-layer PWB thathas been adapted to include a particular stop band filter embodiment. Ascan be seen, the EF_(ext) topology utilized in this example is easilyincorporated into existing PWB designs since it requires neitheradditional metal layers nor any additional thickness. Various otherexternal patch topologies such as the AC_(ext) topology could also beadapted without adding metal layers and the attendant thickness to thePWB. Furthermore, the EFext topology can be implemented as acommensurate structure, such as E2 in FIG. 17, to obtain a broader stopband.

It will be understood by those skilled in the art of panel circuitdesign that the particular type of via embodiment (such as blind,buried, or thru hole vias) employed in fabricating the resonant viaswill vary with the type of panel circuit and other circumstances. Forexample, in PWBs the most common type of via is a plated through-hole(PTH). For this particular embodiment, in may be expedient formanufacturability reasons to construct internal I or internal Tstructures using vias with lengths that extend beyond the internalcapacitive pads, at least to the PPW metal layers. These vias may beconveniently terminated in separate via pads, forming what is known inthe art as a plated thru hole as is shown in FIG. 21. Substitution ofthru hole vias in place of exact length PTH's, such as a blind via in aninternal 1 resonator, has virtually no impact on the fundamental stopband performance of the RF filter since negligible capacitance is addedby extending the via, and negligible inductance is added since thecurrent that travels through the via extension is also negligible.

While certain embodiments of the inventions have been described above,it will be understood that the embodiments described are by way ofexample only. Accordingly, the inventions should not be limited based onthe described embodiments. Rather, the scope of the inventions describedherein should only be limited in light of the claims that follow whentaken in conjunction with the above description and accompanyingdrawings.

1. An electromagnetically reactive structure for attenuating thepropagation of electromagnetic waves comprising: a first conductingplane disposed within a first plane of symmetry in a three-dimensionalperiodic loaded wire media model, a second electrically isolatedconducting plane disposed within a second plane of symmetry in athree-dimensional periodic loaded wire media model, thereby forming aparallel plate waveguide, and a plurality of resonators, each resonatorof the plurality embodying a truncated segment of the three-dimensionalperiodic loaded wire media model and for which some portion thereof isexternal to at least one of the conducting planes for at least someresonators of the plurality.
 2. The electromagnetically reactivestructure of claim 1, wherein the number of resonators in the pluralityof resonators and the location, capacitance, and inductance of eachresonator of the plurality is selected to achieve an electromagneticstop band within the waveguide.
 3. The electromagnetically reactivestructure of claim 2, wherein the electromagnetic stop band is selectedto block transverse propagation of undesirable signals comprisingfrequencies within the stop band.
 4. The electromagnetically reactivestructure of claim 1, wherein the first and second conducting planes aremetallic layers incorporated within a multi-layer performed panelcircuit.
 5. The electromagnetically reactive structure of claim 1,wherein each resonator of the plurality comprises a plated through-holevia.
 6. The electromagnetically reactive structure of claim 5, whereineach resonator of the plurality comprises a first conducting pad coupledwith a plated through-hole via proximate the first end, wherein thefirst pad for at least some resonators of the plurality is in a firstplane that is parallel and external to the first conducting plane. 7.The electromagnetically reactive structure of claim 6, wherein the firstconducting pad for at least some resonators of the plurality is in asecond plane that is parallel with and internal to the first conductingplate.
 8. The electromagnetically reactive structure of claim 7, whereinat least some resonators of the plurality comprise a second conductingpad, and wherein the second conducting pad for at least some resonatorsof the plurality is in a third conducting plane that is parallel withand internal to the second conducting plane.
 9. The electromagneticallyreactive structure of claim 8, wherein the second conducting pad for atleast some resonators of the plurality is in a fourth conducting planethat is parallel with and external to the second conducting plane. 10.The electromagnetically reactive structure of claim 6, wherein the firstpad for at least some resonators of the plurality is in a second planethat is parallel with and external to the second conducting plate.